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04 Sep 2018

04 Sep 2018

# A digital receiver signal strength detector for multi-standard low-IF receivers

Tobias Saalfeld, Tobias Piwczyk, Ralf Wunderlich, and Stefan Heinen

**Tobias Saalfeld et al.**Tobias Saalfeld, Tobias Piwczyk, Ralf Wunderlich, and Stefan Heinen

- RWTH Aachen University, Integrated Analog Circuits and RF Systems Laboratory, 52074 Aachen, Germany

- RWTH Aachen University, Integrated Analog Circuits and RF Systems Laboratory, 52074 Aachen, Germany

**Correspondence**: Tobias Saalfeld (tsaalfeld@ias.rwth-aachen.de)

**Correspondence**: Tobias Saalfeld (tsaalfeld@ias.rwth-aachen.de)

Received: 25 Jan 2018 – Accepted: 25 May 2018 – Published: 04 Sep 2018

This paper presents a receiver signal strength detector based on a discrete Fourier transform implementation. The energy detection algorithm has been designed and measured using a custom multi-standard transceiver ASIC with a low-IF receiver at 0.5, 1 and 2 MHz IF. The proposed implementation directly processes the single bit ΔΣ modulator data and features a clear channel assessment for arbitrary modulation schemes without energy consuming demodulation. Continuous monitoring of the derivative of the RSSI takes advantage of faster coefficient convergence for higher power levels and reduces computation time. A dynamic range of 65 dB has been achieved in FPGA based measurements with a linearity error of less than 1.2 dB. Furthermore, synthesis results for an on-chip implementation for an 130 nm RF CMOS technology show an overall power consumption of 1.5 mW during calculation.

Short range wireless systems with low power consumption are increasingly used for communication of sensor networks. The communication of these networks usually takes place on three bands, which are also used for a various range of common standards. WiFi, Bluetooth or Bluetooth LE are located in the 2.4 GHz band or IEEE 802.15.4 (2016), which specifies multiple physical layers not only for 2.4 GHz, but also for 868 MHz and even for the 433 MHz ISM band. In the last decade all these bands have been increasingly occupied by various communications. Additionally, the trend of the recent years is going towards multi-mode, multi-standard transceivers which offer a wide range of functionality.

With the increased unlicensed use of these bands by various standards, listen
before talk functionality is getting more important to ensure a reliable
communication. These *Clear Channel Assessments* (CCA) can be run
based on the energy detected in a channel. For standards like
IEEE 802.15.4 (2016) a receiver energy detection (ED) is mandatory and a
corresponding value between `0x00`

and `0xFF`

has to be
provided to the user. Furthermore, an accuracy of at least ±6 dB
is required to fulfill the specification.

The low-IF receiver (RX) architecture with ΔΣ modulators (ΔΣM) is a suitable solution for multi-standard operation due to its high reconfigurability (L. Zhang et al., 2013) and a high dynamic range (DR). The high DR relaxes the requirements of the analog front-end (Ho et al., 2011) by reducing the need for high gain as well as a fine tuning of analog gain. Even coarse gain steps in the RF front-end ensure a sufficient signal level at the analog-digital converter (ADC) input and the sensitivity specifications for a standard like IEEE 802.15.4 (2016) can be fulfilled. Accordingly, the automated gain control (AGC) loop of the RX front-end alone does not need a high resolution and can easily be controlled from digital signal processing. One method to determine the receiver signal strength indicator (RSSI) value with a high resolution can be found in the digital AGC loop of the digital baseband (BB) filters. However, these BB filters have a comparatively high current consumption during operation and the digital AGC has a build in hysteresis.

Common single band transceiver architectures like the one presented by Raja et al. (2010) or even the Atmel AT86RF233 (2014) use limiting amplifiers (LA) in the RX front-end to limit the signal amplitude to the DR of the ADC. LA offer the possibility of extracting the RSSI with a rather low additional effort as shown by Huang et al. (2000), but the reception of amplitude modulated signals is not possible in this case. However, in low-IF RX architectures like the one presented by Zolfaghari et al. (2017), LAs are not needed. Consequently, an energy and area efficient solution for the ED, instead of limiting amplifiers, is necessary.

This work presents a digital implementation for RSSI calculation based on a discrete Fourier transform algorithm without area consuming digital multipliers. Furthermore, the RSSI quality is monitored during RSSI calculation to minimize the computation time. The algorithm directly processes the oversampled ΔΣM single bit output data without previous digital filtering. Since the Fourier transform is a highly selective filter itself, down conversion, filtering, demodulation and decimation of the digital BB processing can be deactivated, if the received data is dispensable. This is the case in CCA-ED operation where it has to be ensured that the desired channel is not occupied. Hence, listen before talk functionality can be implemented in an energy efficient way by deactivating digital system components with high power consumption.

The presented RSSI calculation has been designed and tested using synthesized VHDL code executed on an FPGA and connected to a custom triple-band multi-standard transceiver ASIC. An overview of the used RX front-end, the requirements for energy detection and the implementation will be given in the following sections. Subsequently, the RSSI calculation algorithm will be discussed in detail and solutions for an area efficient implementation are given. Furthermore, the measurement setup is presented and the results are shown and compared to analog implementations of ED.

An overview of the relevant parts of the transceiver ASIC is shown as a block diagram of the RX architecture in Fig. 1. The front-end features a sliding-IF mixer for down conversion of the 2.4 GHz band to an intermediate frequency (IF) of 0.5 to 2 MHz depending on the channel bandwidth. The 1.6 GHz local oscillator (LO) signal of the frequency synthesizer is therefore divided by two and used in the second mixer stage. Furthermore the divided LO signal can be connected to the 868 MHz mixers LO input or further divided by two for the 433 MHz mixer. In this way three bands can be supported with only a single phase-locked loop, which is additionally reused for the ASICs transmitter. After down conversion to the desired IF a complex valued quadrature bandpass filter is used for channel selection. Finally, the received signal is converted to the digital domain using a reconfigurable ΔΣM as presented in Saalfeld et al. (2016).

The Fourier transform is a well known and widely used method to analyze a
signal in frequency domain. In Fig. 2 the fast Fourier
transform (FFT) of simulation data of the single bit mode of the
ΔΣM is shown for two scenarios. In one simulation a
signal is present at 750 kHz. The corresponding frequency bin is marked as
*S*_{k,1}. Another simulation has been performed with an input signal
amplitude below the quantization noise floor. The corresponding frequency bin
has been marked with *S*_{k,0}. It can be observed that for both scenarios
the DC amplitude *S*_{0} is in the same range. In order to determine an RSSI
value using a Fourier transform it has to be taken into account that *S*_{k} is
proportional to the run length. Therefore, *S*_{0} can be used to normalize
*S*_{k} to eliminate this proportionality.

The FFT proves to be an efficient algorithm to calculate the entire frequency
spectrum (Ohm and Lüke, 2007). However, more light-weight implementations are
available for single frequency bins. Implementing a discrete Fourier
transform (DFT) as shown in Eq. (1) for single frequency bins
with fixed values of *k* is a well-fitted solution for signal power
detection. Hence, the absolute value of the complex Fourier-coefficient *S*_{k}
represents the signal power in the *k*th frequency bin. Furthermore, each bin
calculation can be seen as highly selective filtering. Therefore, a narrow
band filter for out-of-channel interferer reduction is not necessary.

## 3.1 Goertzel filter

One approach to the DFT implementation is the Goertzel algorithm. The Goertzel algorithm transfer function is shown in Eq. (2) and is further explained by Oppenheim and Schafer (1999).

Expanding the transfer function with the complex conjugate term, it can be split into an IIR-filter and an FIR-filter as shown in Eq. (3).

Figure 3 shows the corresponding *Z*-transfer function
block diagram. The IIR-filter needs one real multiplication per cycle, while
the remaining complex multiplication of the FIR-filter has to be computed
only once for every frequency bin.

The resulting algorithm works well with CPUs or DSPs, which have a build-in multiplying unit with sufficient word length, because it reduces the clock cycles required for a complex multiplication to those of a single real multiplication. A VLSI implementation does not benefit in the same way from the reduced multiplication effort, since the complex multiplier has to be implemented in hardware. Furthermore, the real multiplication requires considerable word length, since every error induced by quantization is multiplied in the following cycles. The word length in this case is dominated by the word length of the constant cos-factor, which should be larger than the input word length to achieve the desired resolution.

## 3.2 Discrete Fourier transform

The DFT can also be directly implemented to compute only a single frequency
bin instead of multiple bins as it would be necessary for a spectral range.
In Eq. (1) the DFT calculation for the *k*th frequency bin was
already shown. Figure 4 shows the corresponding
Z-transform block diagram. It consists of a ${e}^{-j\mathrm{2}\mathit{\pi}\frac{k}{N}\cdot n}$
multiplication stage and an integrator. In contrast to the Goertzel algorithm
a time dependency introduced by the index *n* exists for fixed values of the
selected bin *k* and the spectral resolution *N*.

However, the DFT works well with ΔΣM because the high
resolution is achieved with oversampling and noise shaping instead of a high
number of bits. The higher frequency barely has an effect on the VLSI
implementation, while a higher number of bits would directly increase the
physical size of the multiplier. Utilizing the single bit data stream of the
ΔΣM allows to minimize the size of the multiplier in the
input path. Furthermore, the down conversion of the received signal from IF
to DC is typically performed by the demodulator as shown by Y. Zhang et al. (2013).
Hence, the down conversion multiplication with *e*^{−jω} should already
be implemented by the DSP.

The presented architecture, depicted in Fig. 5, uses the single
bit ΔΣM samples. Since the values represent ±1 the
complex multiplier can be implemented as sign selection. A base 2 logarithm
unit is included to allow linear mapping between the calculated RSSI and the
signal power in dBm. Additionally, the common mode coefficient *S*_{0}
is calculated, as shown in Eq. (4).

The absolute value of the relevant frequency bin, $\left|{S}_{k}\right|$, is normalized to $\left|{S}_{\mathrm{0}}\right|$ to have an RSSI value independent of the run length.

As seen in Eq. (5) the normalization is realized by subtracting the logarithmic values, eliminating the need for a divider unit. Continuous calculation allows variable computation time, running as long as necessary, but as short as possible.

## 4.1 DFT core processing

The calculation of the DFT is shown as the core processing part in the left
section in Fig. 5. As stated in Sect. 3.2 the index
value *n* of the complex *e*^{−jω} multiplication shown in
Fig. 4 changes with every cycle. Therefore, a lookup
table (LUT) has to be introduced. If the computation runs continuously, then
the sample count *N* in Eq. (1) is time variant. Furthermore, the
resulting spectral resolution and therefore the width of the detected
frequency bin is dependent on *N* as well. Higher *N* result in narrower
frequency bins and thus increasing the signal to noise ratio (SNR) in this
specific bin. During computation the observed bin is fixed in regard to the
absolute frequency and is defined by the factor $\frac{k}{N}$ which is
therefore constant and independent over the run length. Hence the computation
can run continuously and does not necessarily need a fixed duration. This
increases the resolution, allowing to detect signals with low power levels
which would otherwise have been indistinguishable from the noise.
Consequently, longer computation time enables the detection of small signals.

Regarding the LUT values the exponential term in Eq. (1) can be replaced using Euler's formula. As seen in Eq. (6) a cosine value as well as a cosine shifted by a quarter period are required.

The cosine argument is periodic at $n=\frac{N}{k}$, where the largest *N* in
the presented system is set by the maximum ΔΣM clock
frequency *f*_{s}=96 MHz. The smallest necessary *k* is set by the
lowest IF *f*_{IF}=0.5 MHz and is the index of the relevant
frequency bins for RSSI detection. This results in a cosine LUT length of
$M=\frac{\mathrm{96}\times {\mathrm{10}}^{\mathrm{6}}}{\mathrm{0.5}\times {\mathrm{10}}^{\mathrm{6}}}=\mathrm{192}$ for a full period. However,
storing a quarter period is sufficient to reconstruct the entire cosine,
reducing the size of the LUT to 48 entries. Reconstruction is done by either
negating the LUT value or reversing the access. The need to access the values
in reverse as well as in regular order simultaneously requires individual
multiplexer-trees for the sine and cosine values.

The input is a complex value defined by the ΔΣM output
values and has to be interpreted as ${D}_{\mathrm{IF}}=I-jQ$. The complex
multiplier at the input is split into four real multiplications, where *s**i**n*
and *c**o**s* represent the sine and cosine term in Eq. (6).
The resulting signal is interpreted as shown in Eq. (7).
The sign selection of *c**o**s* and *s**i**n* values depending on the input signal
*D*_{IF} is shown in Table 1.

If the input from the ΔΣM is negative and the index *n*
requires the LUT value to be inverted, the resulting value will be positive.
Therefore, the inversion is controlled by the exclusive disjunction and
realized within the following accumulator. Reusing the accumulator adder
requires no additional summation stage for the two's-complement. This
accumulator consists of an adder and a register to store the current complex
Fourier coefficient. An additional pipeline register delays further
processing of the Fourier coefficient until at least one full period of the
exponential term in Eq. (1) is calculated. Consequently,
calculations with incomplete coefficients are prevented.

For the detection of the DC value the coefficient *k* in
Eq. (6) is set to *k*=0 as shown in Eq. (4).
Therefore, the multiplication is not necessary and only an accumulation stage
is needed.

## 4.2 Post-processing

The L1-norm is a sufficient method to be used in an RSSI detection. Shown in
Eq. (8) is the result of an L1-norm of the *S*_{k} signal. The
absolute value is generated by checking the sign and, if necessary, negating
ℜ or ℑ of *S*_{k} during the following summation.

For logarithm calculation the value is, as discussed by Mansour et al. (2014), divided into mantissa and exponent. As shown in Fig. 6 the logarithmic value can be expressed as:

The exponent *e* is calculated using a thermometer encoder, while the
mantissa *m* is generated by shifting the original value *X* by *e* bits.
This is performed by a variable shift register, which requires one clock
cycle per shift. The mantissa is then fed to a LUT with a size of 32 values
that maps *m* to log _{2}(*m*). By using a fixed point format the exponent and
the logarithmic mantissa can be concatenated, yielding the final logarithmic
value. Small variances between concurrent log _{2}(*S*_{k}) values are
attenuated, using a moving average filter with a length of four. A signed
subtracter finally calculates the RSSI value.

The difference of the last two RSSI values is evaluated using an approximate derivative of the RSSI signal. Once the RSSI has converged, the absolute value of this difference drops below a threshold and the RSSI calculation can be stopped. Furthermore, the current slope is given out alongside the RSSI, giving a factor for RSSI quality estimation.

Luo et al. (2013)Huang et al. (2000)Figure 7 shows the RSSI value, the individual frequency
bins as well as the derivation of the RSSI, called *Slope*. Using the
existing ASIC and an FPGA, single bit ΔΣM samples have
been recorded and used for simulations. A high language description of the
hardware was used for the simulations, ensuring early verification of the
algorithms functionality. Figure 7 depicts the development
of the RSSI (${S}_{\mathrm{0}}-\stackrel{\mathrm{\u203e}}{{S}_{k}}$), the DC value *S*_{0} and the *S*_{k} value as
well as the slope shown in percent. The data is exemplary shown for −70 and
−50 dBm input power. It can be seen that the DC value *S*_{0} is
equal for both input powers, while $\stackrel{\mathrm{\u203e}}{{S}_{k}}$ settles to higher values
with higher input power. Furthermore, the decreased settling time for
increased input power values can be observed.

The design functionality has also been verified in measurements using VHDL code synthesized on a Xilinx Virtex 5 board. The design was tested using the existing transceiver ASIC in receive operation as presented in Sect. 2. The RF input signals were supplied by a signal generator. Figure 8 shows the RSSI measurement with a step size of 3 dB. Furthermore, the standard deviation over 10 runs is shown for each measured input signal power.

Additionally, the error of the measured RSSI towards an ideal regression is shown in dB with respect to the input power. It can be seen that for all input power measurements the RSSI error stays below 1.2 dB while the standard deviation increases for smaller input signals due to the decreased SNR.

In order to integrate the ED within the transceiver ASIC a synthesis for a
130 nm RF CMOS technology has been evaluated. The measurement
results and worst case synthesis report for power and area consumption are
used for a comparison to analog LAs in Table 2. The DFT
based implementation shows a decent power consumption *P*_{total}=1.5 mW during calculation with a small area of 39 000 µm^{2}. It has to be pointed out that for the analog LA based implementations
the LA is an essential system block of the used RX architecture unlike in a
low-IF architecture. Furthermore, LA would need a separate ADC in order to
provide a digital RSSI value. Additionally, the integration into a
transceiver ASIC would allow the reuse of the down conversion to BB during
demodulation reducing the power consumption of the proposed implementation by
6 %. In regards to the DR and error the DFT based digital implementation
shows an adequate performance.

A receiver signal strength detection algorithm based on a discrete Fourier
transform algorithm has been presented. It evaluates the single bit output
data stream of a ΔΣ modulator without previous signal
manipulation like digital filtering. Furthermore, the ED has been verified in
measurements using a custom ASIC with a low-IF RX. It features a DR of
65 dB with an error of less than 1.2 dB and provides an
input power dependent run time by evaluating the derivative of the RSSI
signal. The synthesized hardware implementation for a 130 nm RF
CMOS technology showed a small area of 39 000 µm^{2} and power
consumption of less than 1.5 mW. Consequently, an efficient DFT
based implementation for ED has been shown which can directly be used as a
low effort solution for RSSI calculation in low-IF architectures with
ΔΣM.

The raw data for simulation, simulation output data and measurement results are available at Zenodo (https://doi.org/10.5281/zenodo.1260387; Saalfeld and Piwczyk, 2018).

The authors declare that they have no conflict of interest.

This article is part of the special issue “Kleinheubacher Berichte 2017”. It is a result of the Kleinheubacher Tagung 2017, Miltenberg, Germany, 25–27 September 2017.

The authors acknowledge the financial support by the German Federal Ministry
of Education and Research (FKZ 16 ESEO154) and the Electronic Components and
Systems for European Leadership Joint Undertaking under grant agreement No.
737434.

Edited by: Jens Anders

Reviewed by: two anonymous referees

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